a) Field of Invention
This invention relates to a brushless motor drive circuit that uses soft switching signals to produce improved torque ripple correction effects.
b) Background Art
The inventor of the present application has filed a patent application for a brushless motor drive circuit that uses soft switching signals to produce improved torque ripple correction effects. This invention is disclosed in Published Japanese Patent Application No. Hei 1-278145. Firstly, this earlier invention will be described by referring to FIG. 5 of the accompanying drawings.
In FIG. 5, a position detecting means comprising Hall devices Hu, Hv and Hw is juxtaposed with a rotor (not shown) having 2.times.n magnetized poles and supplied with power by a power source. The rotor is juxtaposed with a 3-phase stator (not shown) comprising three drive coils Lu, Lv and Lw arranged for a 3-phase configuration and urged to rotate by this stator. The Hall devices Hu, Hv and Hw detect the rotary position of the rotor comprising a rotor magnet (not shown) having 2.times.n magnetized poles and selectively generates three sinusoidal wave signals Vu, Vv and Vw whose phases are differentiated from one another by 120.degree. as shown in FIG. 2(a) depending on the rotary position of the rotor relative to the stator.
Transistors Q1 through Q6, variable current sources CS6 through CS8 for generating an electric current IO, resistors R3 through R5 equally having a resistance Ro and diodes D1 and D2 constitute a Hall amplifying circuit and, at the same time, a signal synthesizing circuit 44. The variable current sources CS6 through CS8 generate an electric current Io that corresponds to the input voltage applied by an adjuster terminal T1 and can be adjusted to an arbitrarily set level by modifying the input voltage from the adjuster terminal T1.
The signal synthesizing circuit 44 amplifies the output signals Vu, Vv and Vw of the Hall devices-Hu, Hv and Hw and logarithmically compresses them so that the signals have the waveforms reduced flat at and near the inflection points to become somewhat rectangular pulse-like signals and are synthesized to produce three phase differentiated soft switching signals Vu2, Vv2 and Vw2 having a waveform as shown in FIG. 2(b). Differently stated, the output signals Vu, Vv and Vw of the Hall devices Hu, Hv, Hw are amplified by the transistors Q1 through Q6 and the collector outputs of the transistors Q1 and Q4 are synthesized to become a soft switching signal Vu2 while the collector outputs of the transistors Q3 and Q6 are synthetically processed to produce a soft switching signal Vv2 and those of the transistors Q2 and Q5 are synthesized into another soft switching signal Vw2.
The soft switching signals Vu2, Vv2 and Vw2 from the signal synthesizing circuit 44 then pass through respective resistors R7 through R12 and are converted into electric currents by a 3-differential amplifier comprising PNP-type transistors Q31 through Q33, NPN-type transistors Q34 through Q36 and variable current sources CS9 and CS10 and the electric currents are amplified by the same 3-differential amplifier. The output currents Iu2, Iv2 and Iw2 as well as Iu3, Iv3 and Iw3 of the 3-differential amplifier are applied to a predriver PD by way of a mirror circuit comprising transistors Q37 through Q42 and Q43 through Q48.
Then, for instance at phase U, the soft switching signal Vu2 from the amplifying and synthesizing circuit 44 passes through the resistors R7 and R10 and is converted into electric currents by the transistors Q31 and Q34, which amplify the currents, the output current of the collector of the transistor Q31 being fed back to the base (point U1) of the transistor Q34 by a mirror circuit constituted by transistors Q43 and Q52 and resistors R17 and R20, the output current of the collector of the transistor Q34 being, on the other hand, fed circuit constituted by transistors Q37 and Q49 and resistors R13 and R16. The level of the currents fed back to the bases of the transistors Q34 and Q31 is held significantly lower than that of the current Io from the variable current sources CS6, CS7 and CS8.
Similarly at phases V and W, the soft switching signals Vv2 and Vw2 from the amplifying and synthesizing circuit 44 respectively pass through the resistors R8, R11, and R9, R12 and are converted into electric currents by the transistors Q32, Q35 and Q33, Q36, which amplify the currents. The output currents of the collectors of the transistors Q32 and Q33 respectively are fed back to the bases (points V1 and W1) of the transistors Q35 and Q36 by mirror circuits respectively constituted by transistors Q44, Q53 and Q45, Q54 and resistors R18, R19 and R20. The output currents of the collectors of the transistors Q35, Q36 are on the other hand, fed back to the bases (points V2 and W2) of the transistors Q32 and Q33 by mirror circuits respectively constituted by transistors Q38, Q50 and Q39, Q51 and resistors R14, R15 and R16. The level of the currents fed back to the bases of the transistors Q32, Q33, Q35 and Q36 is held significantly lower than that of the current Io from the variable current sources CS6, CS7 and CS8.
With an arrangement as described above, voltages Vsu1, Vsv1 and Vsw1 respectively between the cathode s of the diode D2 and the points U1, V1 and W1 and voltages Vsu2, Vsv2 and Vsw2 respectively between the cathode s of the diode and the points U1, V2 and W2 will be as shown in FIG. 5. The combined diodes D6 and D9, D7 and D10 and D8 and D11 operate as amplitude limiters for limiting the amplitudes of the voltages Vsu1, Vsv1, Vsw1, Vsu2, Vsv2 and Vsw2.
Upon receiving an output signal from the predriver PD, a group of transistors Q55, Q56 and Q57 operates to cause source currents .alpha.Iu2, .alpha.Iv2 and .alpha.Iw2 to flow into the respective drive coils Lu, Lv and Lw, whereas another group of transistors Q58, Q59 and Q60 operates to cause sink currents .alpha.Iu3, .alpha.Iv3 and .alpha.Iw3 to flow out of the respective drive coils Lu, Lv and Lw. The source currents .alpha.Iu2, .alpha.Iv2 and .alpha.Iw2 are obtained by multiplying by a the respective input currents Iu2, Iv2 and Iw2 of the predriver PD by means of the predriver PD and the group of transistors Q55, Q56 and Q57. The sink currents .alpha.Iu3, .alpha.Iv3 and .alpha.Iw3 are obtained by multiplying by the respective input currents Iu3, Iv3 and Iw3 of the predriver PD by means of said predriver PD and the group of transistors Q5a, Q59 and Q60.
Electric current detecting resistor Rs detects any electric currents running through the drive coils Lu, Lv and Lw and converts them to voltages. Any voltage across the resistor Rs is negatively fed back to a current feedback amplifier Ai and compared with a motor speed control signal Vctl at an electric current feedback amplifier Ai and an error voltage representing the difference, if any, between them is utilized to regulate the variable current sources CS9 and CS10 and control the electric currents Ictl coming from them. Thus, the current Ictl is so controlled that the current flowing through the resistor Rs is kept constant as long as the control signal Vctl is kept constant and the constant electric current is supplied to the drive coils Lu, Lv and Lw.
FIG. 4 schematically illustrates the waveforms of the source current .alpha.Iu2 and the sink current .alpha.Iu3 for the U-phase drive coil Lu determined by a simulating operation. It may be seen from FIG. 4 that the source current .alpha.Iu2 and the sink current .alpha.Iu3 flow through the current detecting resistor as reactive currents during current conductive periods To when no current flows through the U-phase drive coil Lu as the U-phase transistors Q55 and Q58 are turned on simultaneously, whereas they are never turned on simultaneously during current conductive periods a when a current flows through the U-phase drive coil Lu. A similar statement applies to V- and W-phases.
As described above, there arises a simultaneous on period To when, of the groups of transistors Q55 through Q57 and Q58 through Q60, those having the same phase are turned on and kept on simultaneously if the electric current Io of the variable current sources CS6 through CS8 is set to a low level and a reactive current flows through the current detecting resistor Rs during this simultaneous on period To. Thus, the means for generating simultaneous on periods is constituted by variable current sources CS6 through CS8, transistors for the Hall amplifying circuit Q1 through Q6, resistors R3 through R5 and transistors for the 3-differential amplifier Q31 through Q33 and Q34 through Q36. In the above example no reactive current flows during a current on period a and the means for breaking a reactive current during a current on period is constituted by resistors R7 through R12, diodes D6 through D11, transistors Q49 through Q54 and resistors R10 and R16. Besides, the means for controlling the electric currents supplied to the 3-phase drive coils Lu, Lv and Lw is constituted by current feedback amplifier Ai and variable current sources CS9 and CS10.
Referring to FIG. 4, if any simultaneous on currents flows through transistors having the same phase during a current on period a, those simultaneous on currents will be reactive currents that serve no purpose for driving the motor because those currents are inversely proportional to the amplitude of the soft switching signals Vu2, Vv2 and Vw2. Such reactive currents would interfere with the torque ripple correction effect of a brushless motor and can even aggravate the torque ripples when they deviate from one another.
A circuit as shown in FIG. 5, however, will have below listed effects because reactive currents are broken during any current on period a by the circuit.
(1) The reduction in the magnitude of synthesized torque can be minimized. PA1 (2) Torque ripples can be further improved by reducing the amplitude of the soft switching signals Vu2, Vv2 and Vw2 because no reactive currents flow during current on periods a if they are reduced. PA1 (3) Noises in the motor are lowered when the amplitude of the soft switching signals Vu2, Vv2 and Vw2 is reduced. PA1 (4) Torque ripples are scarcely aggravated if elements of the drive circuits show deviations from standardized performances.
While a brushless motor drive circuit as shown in FIG. 5 shows excellent torque ripple correction effects as described above, there is still room for improvement. For one thing, both the electric conductivity of the power transistors on-the source side and that of the power transistors on the sink side are controlled by a single electric current Ictl to switch the currents supplied to the drive coils and therefore, if the source side circuit and the sink side circuit have unbalanced current gains, the source and sink currents of the drive coils may differ. This will produce a saturated condition in the final stage of the circuits at the side having a higher gain, thereby distorting the waveforms of the electric currents flowing through the coils to degrade their torque ripple correction capability and consequently generating torque ripples to make the rotor of the motor rotate unevenly. For another thing, the voltage at the middle point of each of the coils can become unstable so as to generate torque ripples and, in some cases, cause the circuit to generate oscillation if the source side and the sink side of the drive circuit show a reversed relationship with regard to the magnitude of current gain of the drive circuit for each phase.
It is therefore the object of the present invention to resolve the above mentioned problems of the prior art and provide a brushless motor drive circuit having an improved ripple correction capability that can effectively maintain the voltage of the middle points of the coils to a desired level, or approximately half of the power source voltage, even if the current gain of the source side circuit section and that of the sink side circuit section are not balanced. This will avoid the final stage of both circuit sections of the drive circuit from becoming saturated and thereby generating undesired torque ripples and oscillations.